Power amplifier architectures

ABSTRACT

Implementations and examples of power amplifier devices, systems and techniques for amplifying RF signals, including power amplifier systems based on Composite Right and Left Handed (CRLH) metamaterial (MTM) structures.

PRIORITY CLAIM AND RELATED APPLICATION

This application is a divisional application of U.S. patent applicationSer. No. 12/474,270, entitled, “POWER AMPLIFIER ARCHITECTURES,” andfiled on May 28, 2009, now U.S. Pat. No. 8,180,303 which claims thebenefits of U.S. Provisional Patent Application Ser. No. 61/056,826,entitled, “POWER AMPLIFIER ARCHITECTURES FOR HIGH EFFICIENCY AND HIGHLINEARITY USING METAMATERIAL STRUCTURES,” and filed on May 28, 2008, thespecifications of which are incorporated by reference as part of thespecification of this application.

BACKGROUND

This patent document relates to power amplifier devices, systems andtechniques for amplifying RF signals.

Power amplifiers (PAs) are devices that amplify power of RF signals byconverting DC power into RF power and are used in various RF devices andsystems. Many power amplifiers used in a transceiver system tend tooccupy a large chip or board area, require high DC power, and need todissipate heat generated during operation. Good efficiency and linearityof the PAs are demanded in wireless communications and other RFapplications. Many wireless standards targeting high communicationthroughput utilize complex modulation schemes to generate an RF signalwith both amplitude and phase components. Such schemes may imposestringent linearity requirements on the PAs in the system. In addition,the PAs are required to operate at adequate levels of efficiencies tominimize battery power consumption and heat dissipation.

SUMMARY

Implementations and examples of power amplifier devices, systems andtechniques for amplifying RF signals are provided, including poweramplifier systems based on Composite Right and Left Handed (CRLH)metamaterial (MTM) structures.

In one aspect, a power amplifier system for amplifying an RF signal isprovided to include an input port that receives an input RF signal and apower divider including an input transmission line coupled to the inputport to receive the input RF signal and output branch transmission linesthat are coupled to the input transmission line to split the receivedinput RF signal into branch RF signals that are respectively carried bythe output branch transmission lines. This system includes poweramplifiers that are coupled to the output branch transmission lines ofthe power divider, respectively and each power amplifier receives andamplifies a respective branch RF signal to produce an amplified branchRF signal. A power combiner is provided to include input branchtransmission lines that are respectively coupled to the power amplifiersto receive amplified branch RF signals produced by the power amplifiers,and an output transmission line coupled to the input branch transmissionlines to combine the received amplified branch RF signals into acombined output RF signal. At least one of the power divider and thepower combiner includes a Composite Right and Left Handed (CRLH)Metamaterial (MTM) structure.

In another aspect, a device based on a Composite Right and Left Handed(CRLH) Metamaterial (MTM) structure includes a first transmission lineincluding a Composite Right and Left Handed (CRLH) Metamaterial (MTM)structure, a first terminal that is terminated at a first impedance anda second terminal; and second transmission lines of CRLH MTM structuresthat are coupled to the second terminal of the first transmission lineto either split an input RF signal received from the second terminal ofthe first transmission line into output RF signals in the secondtransmission lines, respectively, or combine input RF signals in thesecond transmission lines into an output RF signal at the secondterminal of the first transmission line. Each second transmission linehas a first terminal that is coupled to the second terminal of the firsttransmission line and a second terminal that is terminated at arespective second impedance. The first transmission line and the secondtransmission lines are structured to provide impedance match between thefirst impedance at the first terminal of the first transmission line anda respective second impedance of each of the second transmission lines.In this device, the first transmission line and the second transmissionlines are structured to be a first configuration or a secondconfiguration. In the first configuration, the first transmission lineis a CRLH MTM transmission line of an electrical length having a phasedelay of ±m90° where m is a positive odd integer, and each of the secondtransmission lines is a CRLH MTM transmission line of an electricallength having a zero phase delay or a phase delay of ±n180° where n is apositive integer. In the second configuration, the first transmissionline is a CRLH MTM transmission line of an electrical length having azero phase delay or a phase delay of ±m180° where m is a positiveinteger and each of the second transmission lines is a CRLH MTMtransmission line having a phase delay of ±n90° where n is a positiveodd integer.

In another aspect, a power amplifier system includes an RF input portfor receiving an input RF signal; an envelope detector coupled to the RFinput port to receive a first portion of the input RF signal to extractinformation on an amplitude envelope of the input RF signal to producean envelope signal; a sigma-delta modulator coupled to the envelopedetector to convert the envelope signal into a first signal; a limitercircuit coupled to the RF input port to receive a second portion of theinput RF signal to eliminate amplitude modulation components of theinput RF signal to produce a second signal that retains phase of theinput RF signal; a time delay circuit coupled to the limiter circuit tocause a delay in the second amplifier input signal to synchronize withthe first signal; a modulator coupled to receive the first and secondsignals and to combine the received first and second signals into athird signal; and a power amplifier coupled to the modulator to receiveand amplify the third signal into an output RF signal which isproportional to the input RF signal. The power amplifier is a CompositeRight and Left Handed (CRLH) Metamaterial (MTM) structure poweramplifier.

In another aspect, a power amplifier system is provided to include an RFinput port for receiving an input RF signal; an envelope detectorcoupled to the RF input port to receive a first portion of the input RFsignal to extract information on an amplitude envelope of the input RFsignal to produce an envelope signal; a sigma-delta modulator coupled tothe envelope detector to convert the envelope signal into a firstsignal; a limiter circuit coupled to the RF input port to receive asecond portion of the input RF signal to eliminate amplitude modulationcomponents of the input RF signal to produce a second signal thatretains phase of the input RF signal; a time delay circuit coupled tothe limiter circuit to cause a delay in the second amplifier inputsignal to synchronize with the first signal; a power amplifier coupledto receive the first and second signals to produce an output RF signalwhich is proportional to the input RF signal; and a filter coupled tothe power amplifier to filter the output RF signal and having aComposite Right and Left Handed (CRLH) Metamaterial (MTM) structure.

In another aspect, a power amplifier system for amplifying an RF signalincludes an input port that receives an input RF signal; andtransmission lines based on Composite Right and Left Handed (CRLH)Metamaterial (MTM) structures and connected in series to receive theinput RF signal from the input port. Each transmission line has a CRLHMTM structure to produce, at an operating frequency, a zero phase delayor a phase delay of n360° where n is a positive or negative integer andincludes a branch output port that splits a portion of the RF signal asa respective branch RF signal for the transmission line and differenttransmission lines producing different respective branch RF signals.This system includes power amplifiers that are coupled to the branchoutput ports of the transmission lines, respectively, and each poweramplifier receives and amplifies a respective branch RF signal toproduce an amplified branch RF signal. A power combiner based on a CRLHMTM structure is coupled to receive the amplified branch RF signals fromthe power amplifiers and to combine the received amplified branch RFsignals into a combined output RF signal.

In another aspect, a power amplifier system for amplifying an RF signalincludes a Zeroth order resonator power divider having a Composite Rightand Left Handed (CRLH) Metamaterial (MTM) structure that receives aninput RF signal and splits the received input RF signal into branch RFsignals in phase with respective to one another, transistors coupled tothe Zeroth order resonator power divider to receive the branch RFsignals, respectively, and to amplify the received branch RF signals,and a Zeroth order resonating antenna based on a CRLH MTM structure andcoupled to transistors to receive amplified branch RF signals.

In yet another aspect, a power amplifier system is provided to includean RF input for receiving an input RF signal and a first quadraturecoupler based on a Composite Right and Left Handed (CRLH) Metamaterial(MTM) structure. This first quadrature coupler has a first input portcoupled to the RF input to receive the input RF signal, a second inputport that is terminated, a first output port and a second output port.The first quadrature coupler splits the input RF signal into a first RFsignal at the first output port and a second RF signal at the secondoutput port that has a phase shift of 90 degrees with respect to thefirst RF signal. A first power amplifier us coupled to the first outputport to receive and amplify the first RF signal and a second poweramplifier is coupled to the second output port to receive and amplifythe second RF signal. This system includes a second quadrature couplerbased on a CRLH MTM structure that includes a first input port coupledto receive the amplified first RF signal from the first power amplifier,a second input port coupled to receive the amplified second RF signalfrom the second power amplifier, a first output port that is terminatedand a second output to produce an RF output signal. The secondquadrature coupler combines signals received at the first and secondinput ports with a 90-degree phase shift to produce the RF outputsignal.

These and other implementations and their variations are described indetail in the attached drawings, the detailed description and theclaims.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A illustrates an Equivalent Circuit Model for CRLH TL with LumpedElements; FIG. 1B illustrates a Dispersion Diagram of Balanced CRLHMetamaterial;

FIG. 2 illustrates a Multi-band Phase Response of LH, RH and CRLH TL, 0°@ f1 and −360° @ f2;

FIG. 3A shows an example of: a Single Band N-Way MTM PowerCombiner/Splitter that has a zero-degree MTM transmission line coupledto N branch MTM transmission lines with electrical lengths of quarterwavelengths

FIG. 3B shows an example of a Single Band N-Way MTM PowerCombiner/Splitter that has a quarter-wavelength MTM transmission linecoupled to N zero-degree branch MTM transmission lines;

FIG. 4A illustrates a multi-band N-Way MTM Power Combiner/Splitter withone Zero Degree CRLH TL and N Quarter Wavelength CRLH TLs;

FIG. 4B illustrates a multi-band N-Way MTM Power Combiner/Splitter withone Quarter Wavelength CRLH TL and N Zero Degree CRLH TL;

FIG. 5A illustrates a multi-band 2-Way MTM Wilkinson PowerSplitter/Combiner;

FIG. 5B illustrates a multi-band N-Way MTM Wilkinson PowerSplitter/Combiner;

FIG. 6A illustrates a general case of a PA with a Power Splitter and aPower Combiner;

FIG. 6B illustrates an MTM Doherty PA and a 2-Way MTM Power Splitter anda 2-Way MTM Power Combiner; FIG. 6C illustrates an N-Way MTM Doherty PA,an N-Way MTM Power Splitter and an N-Way MTM Power Combiner;

FIG. 7 illustrates an MTM Doherty PA with an MTM Wilkinson andconventional components;

FIG. 8 illustrates an MTM Doherty PA and an MTM Wilkinson;

FIG. 9 illustrates an MTM Inverted Doherty PA and an MTM Wilkinson;

FIG. 10 illustrates an MTM Doherty PA and Two MTM Wilkinsons;

FIG. 11A illustrates an Envelope Elimination and Restoration (EER)architecture;

FIGS. 11B-11D illustrates an Integration of MTM PA in EnvelopeElimination and Restoration (EER);

FIG. 12 illustrates a Series Fed Parallel Combining DistributedAmplifier with Zero Degree CRLH TL;

FIG. 13 illustrates a Series Fed Parallel Combining DistributedAmplifier with Zeroth Order Resonator;

FIG. 14 illustrates an N-Port Multi-band Zero Degree CRLH TL;

FIG. 15 illustrates an N-port Multi-band zeroth order resonator;

FIG. 16 illustrates an MTM Amplifying Antenna;

FIG. 17 illustrates an MTM Balanced Power Amplifier;

FIG. 18 illustrates a Multi-band MTM Branch Line Coupler;

FIG. 19 illustrates a Dispersion Diagram of CRLH-TL;

FIG. 20A illustrates a Broadband MTM Branch Line Coupler;

FIG. 20B illustrates a plot of the return and insertion loss comparingthe Broadband MTM Branch Line Coupler with a conventional branch linecoupler; FIG. 20C illustrates a plot of the phase difference between|S21| and |S31|;

FIG. 21 illustrates a Planar MTM Forward-Wave (FW) Directional Coupler;

FIG. 22A illustrates a Vertical MTM Forward-Wave (FW) DirectionalCoupler;

FIG. 22B illustrates a cross-section view of the Vertical MTM FWDirectional Coupler shown in FIG. 22A;

FIG. 23 illustrates an MTM LINC PA; and

FIG. 24 illustrates an MTM LINC PA for different signals.

DETAILED DESCRIPTION

Various power amplifier architectures are available for designing RFdevices and systems. Some implementations for the techniques, devicesand systems described in this document can incorporate features invarious power amplifier architectures.

Linear PAs biased Class AB are widely used in RF applications whileother PA architectures are also used such as certain PA architecturesused in Global System for Mobile communications (GSM), General PacketRadio Service (GPRS) communication systems and Enhanced Data rates forGSM Evolution (EDGE) applications, some of which use the GaussianMinimum Shift Keying (GMSK) modulation. In general, a Class A PA can beused to achieve good linearity at the expense of high DC powerdissipation, thus achieving low efficiency. Class AB PAs can be used toobtain a good compromise of efficiency and linearity. By way ofcomparison, Class AB amplifiers can achieve higher efficiency than ClassA amplifiers.

Many power amplifiers are designed to achieve their maximum efficiencieswhen operated to produce their respective maximum power levels. Variouswireless standards targeting high communication throughput utilizecomplex modulation schemes by manipulating both amplitude and phasecomponents to generate an RF signal. Such techniques may imposestringent linearity requirements on the PAs in the system. At the sametime, the PAs may be required to operate at adequate levels ofefficiencies to minimize battery power consumption and to reduce heatdissipation. Thus, PA designs that achieve good efficiency and linearityare desired for wireless communication systems. In addition to operatingin the linear region while maintaining adequate efficiency, modernwireless applications may require devices to support multiple frequencybands. In some implementations, PAs are designed to operate in theirrespective single bands and are packaged into a single module to providemultiple band operations.

Some modulation schemes used in wireless systems, such as QuadratureAmplitude Modulation (QAM) with Orthogonal Frequency Divisional MultipleAccess (OFDMA) or Code Division Multiple Access (CDMA), create RFsignals that have a high Peak to Average Power Ratio (PAPR) in the rangeof 7-15 dB, for example. To avoid this, the PA is set to operate so asto output the power that is lower than its peak output power until thelinearity requirements are just met. This operation is referred to as“power back-off” and can significantly reduce the efficiency of the PAwhen the PA is designed to achieve its maximum efficiency at or near itsmaximum power level. Typical Power Added Efficiency (PAE) values forOFDM signals (802.11a/g, n, WiMax) are 15% to 20% @ 3% RMS EVM or <10% @2% RMS EVM, where EVM stands for Error Vector Magnitude. In an RF poweramplifier, PAE is defined as the ratio of the difference between theoutput and input RF power to the DC power consumed, whereas drainefficiency is defined as the ratio of the output power to the DC powerconsumed.

In many PA designs, the linearity and good efficiency do not occursimultaneously. The PA output power may be backed off from its maximumpower to meet linearity requirements while compromising the PAefficiency during the backing-off operation.

To increase the output power of a PA, either the voltage swing or thecurrent swing of the PA needs to be increased. Conventionally, the sizeof the device periphery is increased to allow a large current swing. Anincrease in the device size can increase its maximum power, but alsodecreases its optimum load impedance, which is difficult to match. Thearchitectures presented in this application can use any technologyavailable. Examples include printing the designs on a conventional FR-4PCB, Duroid and using thin/thick-film ceramic techniques based onalumina or Low Temperature Co-fired Ceramic (LTCC). Another example is amonolithic fabrication. The architectures presented herein areapplicable to most Monolithic Microwave Integrated Circuits (MMIC)semiconductor processes. These include GaAs-, InGaP-, SiGe-, GaN, InP,Si-based fabrication processes and others. The transistors in thedifferent architectures can be FET, pHEMT, HBT, CMOS, LDMOS and BiCMOStransistors and the like based on the appropriate device processtechnology.

Examples of power amplifier (PA) architectures designed to enhanceefficiency and linearity include the Doherty amplifier, the EnvelopeElimination and Restoration (EER), and the Linear amplification usingNon-linear Components (LINC). Distributed amplifier architectures can beused for broadband amplification. These and other different techniquesand architectures can integrate different classes of amplification.Examples include a class A/B/AB/C/D/E/F/F-1/J or a combination thereof.Each class presents some advantages and disadvantages depending on theperformance requirements.

The Doherty architecture uses two or more power amplifiers and operatesin the back-off region in order to maintain good linearity with goodefficiency. The Doherty architecture has the advantage of being easy toimplement, and there is no need for a high-power modulator thatrestricts the bandwidth requirement. A PA system based on the Dohertyarchitecture can include two power amplifiers: a carrier PA or main PAand one peak PA or auxiliary PA. These two power amplifiers are evenlyor unevenly fed with input power and can be designed to operate in anyof the classes mentioned above. Some Doherty PA implementations use aclass-AB PA for the carrier PA and a class-C PA for the peak PA.

In operation of a Doherty PA system, when the input power is low, thepeak PA is OFF, and the carrier PA is ON, acting as a controlled currentsource seeing twice the optimum resistance. In this state, the carrierPA has reached to one half of its maximum current and its maximumvoltage. When the input power increases to a certain level, the peak PAturns ON and acts as a controlled current source, and the carrier PAacts as a controlled voltage source in this case. When the current ofthe peak PA increases, the impedance seen by the carrier PA decreases.Therefore, the output voltage remains constant. When the input powerreaches its maximum, the two PAs experience optimum impedance which canbe equal to the characteristic impedance of the quarter wavelength. Thecurrent is at a maximum at this level. The peak amplifier modulates theload during the signal peak so as to prevent the carrier PA fromsaturating. It is possible to unevenly feed the Doherty PA by applyingmore power to the peak PA than to the carrier PA in order to fullymodulate the load impedance to the value of the optimum impedance athigh power level. This results in more output power than the case inwhich the PAs are driven with the same amount of power. Inverted Dohertyarchitecture improves carrier cell efficiency at low levels when thepeak PA is OFF. In this case, the challenge is to design the outputmatching with the quarter wavelength.

A distributed power amplifier is a structure that can exhibit broadbandcharacteristics in terms of gain, group delay and impedance matching. Abalanced power amplifier splits an input signal into two identical PAsby using a phase quadrature coupler, and then the two outputs of the PAsare recombined by using another phase quadrature coupler. The advantageof using phase quadrature couplers is that any mismatch reflection fromany of the PAs appears in antiphase and does not affect the input oroutput of the system. The balanced configuration can therefore enhancethe stability of the PA and is insensitive to the load mismatch.

In addition to the above and other PA architectures, the techniques,devices and systems described in this document can be implemented basedon different circuit structures. The propagation of electromagneticwaves in most materials obeys the right handed rule for the (E,H,β)vector fields, where E is the electrical field, H is the magnetic field,and β is the wave vector. The phase velocity direction is the same asthe direction of the signal energy propagation (group velocity) and therefractive index is a positive number. Such materials are “right handed”(RH). Most natural materials are RH materials. Artificial materials canalso be RH materials. A metamaterial (MTM) has an artificial structure.When designed with a structural average unit cell size p much smallerthan the wavelength of the electromagnetic energy guided by themetamaterial, the metamaterial can behave like a homogeneous medium tothe guided electromagnetic energy. Unlike RH materials, a metamaterialcan exhibit a negative refractive index with permittivity ε andpermeability μ being simultaneously negative, and the phase velocitydirection is opposite to the direction of the signal energy propagationwhere the relative directions of the (E,H,β) vector fields follow theleft handed rule. Metamaterials that support only a negative index ofrefraction with permittivity ε and permeability p being simultaneouslynegative are pure “left handed” (LH) metamaterials. Many metamaterialsare mixtures of LH metamaterials and RH materials and thus are CompositeRight and Left Handed (CRLH) metamaterials. A CRLH metamaterial canbehave like a LH metamaterial at low frequencies and a RH material athigh frequencies. Designs and properties of various CRLH metamaterialsare described in, for example, Caloz and Itoh, “ElectromagneticMetamaterials: Transmission Line Theory and Microwave Applications,”John Wiley & Sons (2006). CRLH metamaterials and their applications inantennas are described by Tatsuo Itoh in “Invited paper: Prospects forMetamaterials,” Electronics Letters, Vol. 40, No. 16 (August, 2004).

CRLH metamaterials can be structured and engineered to exhibitelectromagnetic properties that are tailored for specific applicationsand can be used in applications where it may be difficult, impracticalor infeasible to use other materials. In addition, CRLH metamaterialsmay be used to develop new applications and to construct new devicesthat may not be possible with RH materials. Examples of MTM antenna,transmission lines, RF power dividers and combiners can be found, e.g.,in U.S. patent application Ser. No. 11/741,674 entitled “Antennas,Devices and Systems Based on Metamaterial Structures” and filed on Apr.27, 2007 (U.S. Patent Publication No. US20080258981), Ser. No.11/844,982 entitled “Antennas Based on Metamaterial Structures” andfiled on Aug. 24, 2007 (U.S. Patent Publication No. US20080048917), Ser.No. 11/963,710 entitled “Power Combiners and Dividers Based on CompositeRight and Left Handed Metamaterial Structures” and filed on Dec. 21,2007 (U.S. Patent Publication No. 2009/0160575), and Ser. No. 12/340,657entitled “Multi-Metamaterial-Antenna Systems with Directional Couplers”and filed on Dec. 20, 2008 (U.S. Patent Publication No. 2009/0219213),which are incorporated by reference as part of the disclosure of thisdocument. The circuit structures for the present PA architectures can bebased on RH material structures or CRLH MTM structures. As an example,the transmission lines used in the present PA architectures can beimplemented by CRLH Metamaterial (MTM) transmission lines or circuitcomponents to achieve multi-band or wide-band operations in compact formfactors while complying with impedance and phase matching conditions.Consider a simple example of a 1-dimensional structure of CRLH MTMelements which can include MTM unit cells characterized by the cellshunt inductance (LL), the cell shunt capacitance (CR), the cell seriesinductance (LR) and the cell series capacitance (CL). These parametersand the number of MTM unit cells determine the resonant frequencies, thecorresponding bandwidth, input and output impedance matching and thecircuit structure and size. The LH properties of MTM structures allowthe dimensions of MTM structures to be smaller than equivalent circuitsmade of RH materials, e.g., a fraction of λ/2, where λ is thepropagation wavelength of an RF signal in the free-space. An CRLHtransmission line can be designed to have an electrical length thatcorresponds to various phase values, including a phase of zero degree,180 degrees or a multiple of 180 degrees, 90 degrees or a multiple of 90degrees at an operating signal frequency. A CRLH structure can also bedesigned to support two or more frequency bands or broad bandoperations. CRLH MTM structures can be used in PA systems to providemulti-band operation, high efficiency, variable output power, andcompact form factor while complying with matching conditions. Examplesin this document include (1) power amplifier designs multiple PA systemssuch as Doherty PA architectures using the MTM technology in variousconfigurations such as single band MTM PA systems, multi-band MTM PAsystems, MTM Wilkinson or MTM power combiners and splitters, and MTMInverted Doherty PA; (2) envelope elimination restoration (EER) PAsystems including EER PAs that drive drain bias of an MTM PA and mixdigital signals before feeding an MTM PA; (3) MTM balanced PA systems inmulti-band and broadband configurations; and (4) MTM LinearAmplification using Non-linear Components (LINO).

A pure Left Handed (LH) material follows the left hand rule for thevector trio (E,H,β), and the phase velocity direction is opposite to thesignal energy propagation. Both the permittivity and permeability of theLH material are negative. A Composite Right/Left Handed (CRLH)metamaterial can exhibit both left hand and right hand electromagneticmodes of propagation depending on the regime or frequency of operation.Under certain circumstances, a CRLH metamaterial can exhibit a non-zerogroup velocity when the wavevector of a signal is zero. This situationoccurs when both left hand and right hand modes are balanced. In anunbalanced mode, there is a bandgap in which electromagnetic wavepropagation is forbidden. In the balanced case, the dispersion curvedoes not show any discontinuity at the transition point of thepropagation constant β (ω_(o))=0 between the Left and Right handedmodes, where the guided wavelength is infinite λ_(g)=2π/|β|→∞ while thegroup velocity is positive:

$\begin{matrix}{v_{g} = {\frac{\mathbb{d}\omega}{\mathbb{d}\beta}❘_{\beta = 0}{> 0}}} & (1)\end{matrix}$This state corresponds to the Zeroth Order mode m=0 in a TransmissionLine (TL) implementation in the LH region. The CRHL structure supports afine spectrum of low frequencies with a dispersion relation that followsthe negative β parabolic region which allows a physically small deviceto be built that is electromagnetically large with unique capabilitiesin manipulating and controlling near-field radiation patterns. When thisTL is used as a Zeroth Order Resonator (ZOR), it allows a constantamplitude and phase resonance across the entire resonator. The ZOR modemay be used to build MTM-based power combiners and splitters ordividers, directional couplers, matching networks, and leaky waveantennas. Examples of MTM-based power combiners and dividers aredescribed below.

In RH TL resonators, the resonance frequency corresponds to electricallengths θ_(n)=β_(n)l=mπ (m=1, 2, 3, . . . ), where l is the length ofthe TL. The TL length should be long enough to reach low and widespectrum of resonant frequencies. The operating frequencies of a pure LHmaterial are at low frequencies. A CRLH metamaterial structure is verydifferent from RH and LH materials and can be used to reach both highand low spectral regions of the RF spectral ranges of RH and LHmaterials. In the CRLH case, θ_(m)=β_(n)l=mπ, where l is the length ofthe CRLH TL and the parameter m=0, ±1, ±2, ±3, . . . , ±∞.

FIG. 1A illustrates an equivalent circuit of an MTM transmission line101 with unit cell 103 in which different unit cells can be connectedtogether in series in a periodic configuration. The equivalent circuitfor each unit cell 103 has a right-handed (RH) series inductance L_(R),a shunt capacitance C_(R) and a left-handed (LH) series capacitanceC_(L), and a shunt inductance L_(L). The shunt inductance L_(L) and theseries capacitance C_(L) are structured and connected to provide theleft handed properties to the unit cell 103. This CRLH TL can beimplemented by using distributed circuit elements, lumped circuitelements or a combination of both. Each unit cell 101 is smaller thanλ/10 where λ is the wavelength of the electromagnetic signal that istransmitted in the CRLH TL. CRLH TLs possess interesting phasecharacteristics such as anti-parallel phase, group velocity, non-linearphase slope and phase offset at zero frequency.

FIG. 1B shows the dispersion diagram of a balanced CRLH metamaterialwith the unit cell in FIG. 1A. The CRLH structure can support a finespectrum of low frequencies and produce higher frequencies including thetransition point with m=0 that corresponds to infinite wavelength. Thiscan be used to provide integration of CRLH antenna elements, such as butnot limited to directional couplers, matching networks, amplifiers,filters, and power combiners and splitters. In some implementations, RFor microwave circuits and devices may be made of a CRLH MTM structure,such as but not limited to directional couplers, matching networks,amplifiers, filters, and power combiners and splitters.

Referring back to FIG. 1A, in the unbalanced case whereL_(R)C_(L)≠L_(L)C_(R), two different resonant frequencies exist: ω_(se)and ω_(sh) that can support an infinite wavelength given by equations:

$\begin{matrix}{\omega_{sh} = \frac{1}{\sqrt{C_{R}L_{L}}}} & (2)\end{matrix}$

$\begin{matrix}{\omega_{se} = {\frac{1}{\sqrt{C_{L}L_{R}}}.}} & (3)\end{matrix}$At ω_(se) and ω_(sh) the group velocity (v_(g)=dω/dβ) is zero and thephase velocity (vp=ω/β) is infinite. When the series and shuntresonances are equal, i.e., L_(R)C_(L)=L_(L)C_(R), the structure is saidto be balanced, and the resonant frequencies coincide:ω_(se)=ω_(sh)=ω₀  (4)

For the balanced case, the phase response can be approximated by thefollowing equations:

$\begin{matrix}{\varphi_{C} = {{\varphi_{RH} + \varphi_{LH}} = {{{- \beta}\; l} = {- \frac{{Nl}\;\omega}{c}}}}} & (5)\end{matrix}$φ_(RH) ≈−N2πf√{square root over (L _(R) C _(R))}  (6)

$\begin{matrix}{\varphi_{LH} \approx \frac{N}{2\pi\; f\sqrt{L_{L}C_{L}}}} & (7)\end{matrix}$where N is the number of unit cells. The slope of the phase is given byequation:

$\begin{matrix}{\frac{\mathbb{d}\varphi_{CRLH}}{\mathbb{d}f} = {{{- {N2}}\;\pi\sqrt{L_{R}C_{R}}} - \frac{N}{2\;\pi\; f^{2}\sqrt{L_{L}C_{L}}}}} & (8)\end{matrix}$The characteristic impedance is given by equation:

$\begin{matrix}{Z_{o}^{CRLH} = {\sqrt{\frac{L_{R}}{C_{R}}} = {\sqrt{\frac{L_{L}}{C_{L}}}.}}} & (10)\end{matrix}$The inductance and capacitance values may be selected and controlled tocreate a desired slope for a chosen frequency. In addition, the phasemay be set to have a positive phase offset at DC. These two factors areused to provide the designs of multi-band and other MTM power combiningand dividing structures presented herein.

The following sections provide examples of determining MTM parameters ofmulti-band mode MTM structures, and similar techniques can be used todetermine MTM parameters with three or more bands.

In a multi-band MTM structure, the signal frequencies f₁, f₂ for the twobands are first selected for two different phase values: φ₁ at f₁ and φ₂at f₂. In this example, let N be the number of unit cells in the CRLH TLand Z_(t), the characteristic impedance. The values for parametersL_(R), C_(R), L_(L) and C_(L) can be calculated as:

$\begin{matrix}{L_{R} = {{\frac{Z_{t}\left\lbrack {{\phi_{1}\left( \frac{\omega_{1}}{\omega_{2}} \right)} - \phi_{2}} \right\rbrack}{N\;{\omega_{2}\left\lbrack {1 - \left( \frac{\omega_{1}}{\omega_{2}} \right)^{2}} \right\rbrack}}\mspace{14mu} C_{R}} = \frac{{\phi_{1}\left( \frac{\omega_{1}}{\omega_{2}} \right)} - \phi_{2}}{N\;\omega_{2}{Z_{t}\left\lbrack {1 - \left( \frac{\omega_{1}}{\omega_{2}} \right)^{2}} \right\rbrack}}}} & (11)\end{matrix}$

$\begin{matrix}{L_{L} = {{\frac{{NZ}_{t}\left\lbrack {1 - \left( \frac{\omega_{1}}{\omega_{2}} \right)^{2}} \right\rbrack}{\omega_{1}\left\lbrack {\phi_{1} - {\left( \frac{\omega_{1}}{\omega_{2}} \right)\phi_{2}}} \right\rbrack}\mspace{14mu} C_{L}} = \frac{N\left\lbrack {1 - \left( \frac{\omega_{1}}{\omega_{2}} \right)^{2}} \right\rbrack}{\omega_{1}{Z_{t}\left\lbrack {\phi_{1} - {\left( \frac{\omega_{1}}{\omega_{2}} \right)\phi_{2}}} \right\rbrack}}}} & (12)\end{matrix}$

$\begin{matrix}{Z_{0}^{CRLH} = {\sqrt{\frac{L_{R}}{C_{R}}} = {\sqrt{\frac{L_{L}}{C_{L}}}.}}} & (13)\end{matrix}$

In the unbalanced case, the propagation constant is given by equation:

$\begin{matrix}{\beta = {{s(\omega)}\sqrt{{\omega^{2}L_{R}C_{R}} + \frac{1}{\omega^{2}L_{L}C_{L}} - \left( {\frac{L_{R}}{L_{L}} + \frac{C_{R}}{C_{L}}} \right)}}} & (14)\end{matrix}$where

$\begin{matrix}{{s(\omega)} = \left\{ \begin{matrix}{- 1} & {if} & {{\omega < {\min\left( {\omega_{se},\omega_{sh}} \right)}}:} & {LH} & {range} \\{+ 1} & {if} & {{\omega > {\max\left( {\omega_{se},\omega_{sh}} \right)}}:} & {RH} & {range}\end{matrix} \right.} & (15)\end{matrix}$

For the balanced case:

$\begin{matrix}{\beta = {{\omega\sqrt{L_{R}C_{R}}} - \frac{1}{\omega\sqrt{L_{L}C_{L}}}}} & (16)\end{matrix}$A CRLH TL has a physical length of d with N unit cells each having alength of p: d=N.p. The signal phase value is φ=−βd. Therefore,

$\begin{matrix}{\beta = {- \frac{\phi}{d}}} & (17)\end{matrix}$

$\begin{matrix}{\beta_{1} = {- \frac{\phi_{i}}{\left( {N.p} \right)}}} & (18)\end{matrix}$

It is possible to select two different phases φ₁ and φ₂ at two differentfrequencies f1 and f2, respectively:

$\begin{matrix}\left\{ \begin{matrix}{\beta_{1} = {{\omega_{1}\sqrt{L_{R}C_{R}}} - \frac{1}{\omega_{1}\sqrt{L_{L}C_{L}}}}} \\{\beta_{2} = {{\omega_{2}\sqrt{L_{R}C_{R}}} - \frac{1}{\omega_{2}\sqrt{L_{L}C_{L}}}}}\end{matrix} \right. & (19)\end{matrix}$

In comparison, a non-MTM RH microstrip transmission line exhibits thefollowing dispersion relationship:

$\begin{matrix}{{\beta_{n} = {\beta_{0} + {\frac{2\pi}{p}n}}},{n = 0},{\pm 1},{\pm 2},\ldots} & (20)\end{matrix}$See, for example, the description on page 370 in Pozar, “MicrowaveEngineering,” 3rd Edition and page 623 in Collin, “Field Theory ofGuided Waves,” Wiley-IEEE Press; 2nd Edition (Dec. 1, 1990).

Multi-band CRLH TL devices can be designed based on a matrix approachdescribed in U.S. patent application Ser. No. 11/844,982. Under thismatrix approach, each 1D CRLH transmission line includes N identicalcells with shunt (L_(L), C_(R)) and series (L_(R), C_(L)) parameters.These five parameters, i.e., (L_(L), C_(R)), (L_(R), C_(L)) and Nidentical cells, determine the N resonant frequencies and phase curves,corresponding bandwidth, and input/output TL impedance variations aroundthese resonances.

The frequency bands are determined from the dispersion equation derivedby letting the N CRLH cell structure resonate with nπ propagation phaselength, where n=0, ±1, . . . ±(N−1). In other words, a zero and 2π phaseresonances can be accomplished with N=3 CRLH cells. Furthermore, atri-band power combiner and splitter can be designed using N=5 CRLHcells where zero, 2π, and 4π cells are used to define resonances.

The n=0 mode resonates at ω₀=ω_(SH), and higher frequencies are given bythe following equation for the different values of M specified in Table1:

$\begin{matrix}{{{{For}\mspace{14mu} n} > 0},{\omega_{\pm n}^{2} = {\frac{\omega_{SH}^{2} + \omega_{SE}^{2} + {M\;\omega_{R}^{2}}}{2} \pm \sqrt{\left( \frac{\omega_{SH}^{2} + \omega_{SE}^{2} + {M\;\omega_{R}^{2}}}{2} \right)^{2} - {\omega_{SH}^{2}\omega_{SE}^{2}}}}}} & (21)\end{matrix}$Table 1 provides M values for N=1, 2, 3, and 4.

TABLE 1 Resonances for N = 1, 2, 3 and 4 cells N \Modes |n| = 0 |n| = 1|n| = 2 |n| = 3 N = 1 M = 0; ω₀ = ω_(SH) N = 2 M = 0; ω₀ = ω_(SH) M = 2N = 3 M = 0; ω₀ = ω_(SH) M = 1 M = 3 N = 4 M = 0; ω₀ = ω_(SH) M = 2 − √2M = 2

FIG. 2 shows an example of the phase response of a CRLH TL which is acombination of the phase of the RH components and the phase of the LHcomponents. Phase curves for CRLH, RH and LH transmission lines areshown. The CRLH phase curve approaches the LH TL phase at lowfrequencies and approaches the RH TL phase at high frequencies. Notably,the CRLH phase curve crosses the zero-phase axis with a frequency offsetfrom zero. This offset from zero frequency enables the CRLH curve to beengineered to intercept a desired pair of phases at any arbitrary pairof frequencies. The inductance and capacitance values of the LH and RHcan be selected and controlled to create a desired slope with a positiveoffset at the zero frequency (DC). By way of example, FIG. 2 shows thatthe phase chosen at the first frequency f₁ is 0 degree, and the phasechosen at the second frequency f₂ is −360 degrees. In addition, a CRLHTL can be used to obtain an equivalent phase with a much smallerfootprint than an RH transmission line.

Therefore, CRLH power combiners and dividers can be designed forcombining and dividing signals at two or more different frequenciesunder impedance matched conditions to achieve compact devices that aresmaller than conventional combiners and dividers. Referring back to FIG.1A, each CRLH unit cell 103 can be designed based on different unitconfigurations in CRLH power combiners and dividers. The properties ofthe metamaterial offer new possibilities for different types of designfor dual-frequencies but also for multi-band systems.

One example of a power amplifier system for amplifying an RF signalbased on a CRLH MTM structure can be designed to include a powerdivider, multiple power amplifiers connected in parallel and a powercombiner. Both or just one of the power divider and the power combinercan be a CRLH MTM structure to achieve certain technical features. Inthis system, an input port is provided to receive an input RF signal.The power divider includes (1) an input CRLH transmission line coupledto the input port to receive the input RF signal and (2) multiple outputCRLH transmission lines that are coupled to the input CRLH transmissionline to split the received input RF signal into different branches. Thepower amplifiers are coupled to the output branch transmission CRLHlines of the power divider, respectively. Each power amplifier receivesand amplifies an RF signal to produce an amplified RF signal. The powercombiner in this system includes multiple input CRLH transmission linesthat are respectively coupled to the power amplifiers to receiveamplified RF signals produced by the power amplifiers. Each input CRLHtransmission line of the power combiner is connected to one poweramplifier. This combiner also includes an output CRLH transmission linecoupled to the input CRLH transmission lines to combine the receivedamplified branch RF signals into a combined output RF signal.

The above MTM-based multi-PA system can be implemented as a MTM-enhancedDoherty PA system. The Doherty technique may achieve high efficiency inthe back-off region. Also, by using the property of the MTM, linearitymay be enhanced. A non-MTM Doherty PA has a large size due to presenceof quarter wavelength transformers designed on microstrip which supporta single-band. MTM-enhanced Doherty PA can replace the single-bandquarter wavelength transformers with multi-band MTM quarter wavelengthtransformers which are also more compact. The power splitter at theinput can be implemented by a compact MTM multi-band power splitter. Theinput/output matching networks may be integrated in the power combinerfor better efficiency. An N-way power combiner/splitter may be also usedwith each branch matched to the desired impedance. The use of MTMelements may increase the bandwidth of operation by being able tocontrol the phase slope. In addition, the MTM properties may facilitatethe connection of antennas with poor Voltage Standing Wave Ratio (VSWR),for example.

The power divider or combiner in the above MTM-based multi-PA system canbe designed to include a first transmission line including a CRLH MTMstructure and multiple second transmission lines of CRLH MTM structuresthat are coupled to one terminal of the first transmission line toeither split an input RF signal received from the first transmissionline into multiple output RF signals in the second transmission lines,respectively, when configured as a power divider, or, when configured asa power combiner, combine input RF signals in the second transmissionlines into an output RF signal in the first transmission line. The firsttransmission line and the second transmission lines are structured to bein two configurations. In the first configuration, the firsttransmission line is a CRLH MTM transmission line of an electricallength having a phase delay of ±m90° where m is a positive integer, andeach of the second transmission lines is a CRLH MTM transmission line ofan electrical length having a zero phase delay or a phase delay of±n180° where n is a positive integer. In the second configuration, thefirst transmission line is a CRLH MTM transmission line of an electricallength having a zero phase delay or a phase delay of ±m180° where m is apositive integer and each of the second transmission lines is a CRLH MTMtransmission line having a phase delay of ±n90° where n is a positiveinteger. The transmission lines can be designed to have a single RFfrequency or two or more different RF frequencies.

FIGS. 3A and 3B show examples of two different single band N-way MTMpower combiner/splitters. These two MTM power combiner/splitters can bedesigned to achieve a more compact package with respect to non-MTM powercombiner/splitters.

In one implementation shown in FIG. 3A, each of the quarter wavelength(φ₁₂) CRLH-TLs 301 are connected at one end to its correspondingimpedance 303, and at the other end to a zero degree (φ₁₁) CRLH-TL 305.The phase φ_(n) is the phase delay caused by the electrical length andis at 0°, ±180°, ±360°, . . . , or a multiple of 180° (i.e., n180° wheren is a positive or negative integer), and the phase φ₁₂ is the phasedelay caused by the electrical length and is at ±90°, ±270°, . . . or amultiple of 90° (n90° where n is a positive or negative odd integer). Inthis application, the “quarter wavelength” refers to an odd multiple ofa quarter wavelength, i.e. (2n+1)×λ/4, where n=1, 2, 3 . . . . Asillustrated, the quarter wavelength (φ₁₂) CRLH-TLs 301 terminated attheir respective impedances Z₁, Z₂, . . . and Z_(N).

The quarter wavelength (φ₁₂) CRLH-TLs 301 and the zero-degree (φ₁₁)CRLH-TL 305 are designed to provide impedance matching between each ofthe impedances Z₁, Z₂, . . . and Z_(N) to the impedance 307 (which canbe, e.g., 50 Ohms as shown) at the terminal end of the zero-degree (φ₁₁)CRLH-TL 305. In this regard, each of the quarter wavelength (φ₁₂)CRLH-TLs is used to transform the corresponding impedance to match theimpedance 307 (e.g., 50 ohms as shown). The number of ports, N, may beselected based on the requirement of a specific application. In theimplementation shown in FIG. 3A, the zero degree (φ₁₁) CRLH-TL 305 isconnected between the impedance 307 at one terminal end of the CRLH-TL305 and the quarter wavelength (φ₁₂) CRLH-TLs 301 at the other terminalend of the CRLH-TL 305. In another implementation, the quarterwavelength (φ₁₂) CRLH-TLs 301 may be connected directly to the 50-ohmimpedance 307 without the zero degree (φ₁₁) CRLH-TL 305. For the exampleof the impedance 307 at 50 ohm, the impedance of each quarter wavelengthCRLH-TL 301 is determined byZ ₀=√{square root over ((50*Z _(J) *N))},where J=1, . . . , or N, in either configuration.

FIG. 3B shows another implementation in which each of the zero degree(φ₁₁) CRLH-TLs 321 is connected at one end to an impedance Z₁ 323 and atthe other end to a quarter wavelength (φ₁₂) CRLH-TL 325. N number of theimpedances with the same value Z₁ 323 connected in parallel aretransformed to 50 ohms 327 by the quarter wavelength (φ₁₂) CRLH-TL 325.For the example of the impedance 327 at 50 ohm, the impedance of thequarter wavelength (φ₁₂) CRLH-TL 325 is determined by

$Z_{0} = \sqrt{\left( {50*Z_{1}*\frac{1}{N}} \right)}$

FIGS. 4A and 4B presents two examples of multi-band N-way MTM powercombiner/splitters which may be used to combine or split the power inthe Doherty architecture or other PA architectures.

In the implementation shown in FIG. 4A, each of quarter wavelength (φ₁₂,φ₂₂) CRLH-TLs 401 may be connected at one end to its correspondingimpedance 403, i.e. one of Z₁, Z₂, . . . and Z_(N), and at the other endto a zero degree (φ₁₁, φ₂₁) CRLH-TL 405, where both φ₁₁ and φ₂₁ are setto be any one of 0°, ±180°, ±360°, . . . ±n180° (n is a positive ornegative integer). Both φ₁₂ and φ₂₂ are set to be any of ±90°, ±270°, .. . ±(2n+1)90° where n is a positive or negative integer and (2n+1) isan odd integer. At one frequency f₁, each quarter wavelength (φ₁₂, φ₂₂)CRLH-TL 401 provides a phase of φ₁₂, and at another frequency f₂ itprovides a phase of φ₂₂. The zero degree (φ₁₁, φ₂₁) CRLH-TL 405 has aphase of φ₁₁ at the frequency f₁ and a phase of φ₂₁ at the otherfrequency f₂. Each of the quarter wavelength (φ₁₂, φ₂₂) CRLH-TLs 401 isused to transform the corresponding impedance to 50 ohms 407, where thenumber of ports, N, can be arbitrary. The impedance of each quarterwavelength CRLH-TL 401 is determined by Z₀=√{square root over((50*Z_(j)*N))}, j=1, . . . , or N, for both φ₁₂ and φ₂₂.

In another implementation shown in FIG. 4B, each of zero degree (φ₁₁,φ₂₁) CRLH-TLs 421 may be connected at one end to an impedance Z₁ 423 andat the other end to a quarter wavelength (φ₁₂, φ₂₂) CRLH-TL 425. Nnumber of the impedances with the same value Z₁ 423 connected inparallel are transformed to 50 ohms 427 by the quarter wavelength (φ₁₂,φ₂₂) CRLH-TL 425. The impedance of the quarter wavelength (φ₁₂, φ₂₂)CRLH-TL 425 is determined by

$Z_{0} = \sqrt{\left( {50*Z_{1}*\frac{1}{N}} \right)}$for both φ₁₂ and φ₂₂.

In yet another implementation, the series capacitor CL in the equivalentcircuit shown in FIG. 1A may be replaced with a varactor. The bias ofthe varactor is adjusted in order to control the impedance matching ofthe CRLH-TL to compensate for the impedance change depending on thenumber of multiple CRLH-TLs that are operating in the “ON” state.

Wilkinson power combiners and splitter can also be designed based on MTMstructures. A Wilkinson power splitter, for example, can include a firstCRLH MTM transmission line as the input transmission line of the powersplitter, a second CRLH MTM transmission line as one of the branchoutput transmission lines of the power splitter to have a first terminalcoupled to the first CRLH MTM transmission line and a second terminal. Athird CRLH MTM transmission line is provided as another one of thebranch output transmission lines to have a first terminal coupled to thefirst CRLH MTM transmission line and a second terminal. A resistor iscoupled as a bridge between the second and third CRLH MTM transmissionlines by coupling its first terminal to the second terminal of thesecond CRLH MTM transmission line and by coupling its second terminal tothe second terminal of the third CRLH MTM transmission line. ThisWilkinson power splitter also includes a fourth CRLH MTM transmissionline coupled to the second terminal of the third CRLH MTM transmissionline and a fifth CRLH MTM transmission line coupled to the secondterminal of the third CRLH MTM transmission line. The fourth and fifthtransmission lines provide the outputs of the Wilkinson power splitter.

A Wilkinson power combiner is a reversed design of the power splitter.Such a combiner is placed downstream from power amplifiers to receivethe outputs of the power amplifiers. In the example of two poweramplifiers, the combiner can include a first CRLH MTM transmission linehaving a first terminal coupled to a first power amplifier to receive afirst amplified RF signal and a second terminal to output the firstamplified RF signal, and a second CRLH MTM transmission line having afirst terminal coupled to a second power amplifier to receive a secondamplified RF signal and a second terminal to output the second amplifiedRF signal. A resistor is provided to have its first terminal coupled tothe second terminal of the first CRLH MTM transmission line and itssecond terminal coupled to the second terminal of the second CRLH MTMtransmission line. In addition, a third CRLH MTM transmission line iscoupled to the second terminal of the first CRLH MTM transmission lineand a fourth CRLH MTM transmission line is coupled to the secondterminal of the fourth CRLH MTM transmission line. The combiner furtherincludes a fifth CRLH MTM transmission line forming the outputtransmission line and coupled to the third and fourth CRLH MTMtransmission lines to combine their outputs into a single output.

FIG. 5A shows an example of a 2-Way MTM Wilkinson PowerSplitter/Combiner. In this implementation, a multi-band input RF signalis received at an input port 501 having frequencies f₁ and f₂, forexample. The RF input signal may be unevenly or evenly divided into twobranches. Each CRLH-TLs 503 may be connected at one end to itscorresponding resistor 505, i.e. one of R, and at the other end to aCRLH-TL 507, where the impedance of each CRLH-TL 503 is determined bythe impendence Z_(0,1) from the following equations:k ² =P ₃ /P ₂,where k is the ratio of predetermined uneven powers, P₂ and P₃, atoutputs port 2 and port 3, respectively,Z _(0,3) =Z _(0,1)√(1+k ²)/k ³,Z _(0,3) =Z _(0,1)√{square root over ((1+k ²)/k ³)}Z _(0,2) =K ² Z _(0,3),andR=Z _(0,1)(k+1/k)The two impedances with the value Z_(0,2) and Z_(0,3) corresponding toeach CRLH-TLs 503 are transformed to 50 ohms 509 by correspondingCRLH-TLs 511.

FIG. 5B illustrates an implementation of an N-Way MTM Wilkinson PowerSplitter/Combiner. In this implementation, a multi-band input RF signalis received at an input port 521 having frequencies f₁ and f₂, forexample. The RF input signal may be unevenly or evenly divided into Nbranches. Each CRLH-TLs 523 can be connected at one end to itscorresponding resistor 525, i.e. one of R₁, R₂, . . . and R_(N), and atthe other end to a CRLH-TL 527, where the impedance of each CRLH-TL 523is determined by the input impendence Z_(0,1). The other end of eachcorresponding resistor 525 is connected to each other to form a singleconnection between all N resistors in this Wilkinson design. In thisexample, the N impedances with the value Z_(0,N) corresponding to eachCRLH-TLs 523 can be transformed to the impedance 529 (which can be,e.g., 50 ohms) by the corresponding CRLH-TLs 531.

Based on the above MTM-based power combiners or splitters, FIG. 6A showsan example of a single or multi-band PA system comprising an N-way powersplitter 601 and an N-way power combiner 603. At least one of thedevices 601 and 603 is based a CRLH MTM design. In operation, the PAsystem amplifies the low input power of the RF signal received at aninput port into higher levels of RF output power transmitted at anoutput port by converting DC power into RF power. For example, an inputsignal is received at the input port 600 coupled to an N-way PowerSplitter 601 where the input signal is unevenly or evenly divided into Nbranches forming multiple output signals. Each output signals of theN-way Power Splitter A 601 is connected to the input of each PA_(n) 605,where n=1, 2, . . . N. The output of each PA_(n) 605 is connected toeach input of an N-way Power Combiner 603, where n=1, 2, . . . N. Theoutput of the N-way Power Combiner 603 transmits the amplified combinedinput signal to an output port 607.

Various implementations of the PA-Power Splitter/Combiner systemdescribed above and shown in FIG. 6A can be made. For example, an MTMpower splitter can be used in FIG. 6A to reduce the size and, thus,integrate the PA system on a microchip depending on the substrate. TheMTM structures can be used to enable multi-band operations. The MTMapplication described for the power splitter can be applied to the powercombiner at the output of the system resulting in similar sizingbenefits and multi-band properties.

The exemplary systems described in this document can utilize PAs such asDoherty PAs, Balanced PAs (e.g., MTM Balanced PAs and non-MTM BalancedPAs), and other MTM and non-MTM PAs. Various types and combinations ofN-way splitters and N-way combiners may be implemented in the PA systemshown in FIG. 6A. For example, each N-way Power Splitter 601 and N-waycombiner 603 may be formed from MTM or conventional structures. Examplesof the various combinations of N-way Power Splitter 601 and N-way PowerCombiner 603 shown in FIG. 6A may include conventional splitters/MTMcombiners, MTM splitters/conventional combiners, and MTM splitters/MTMcombiners. In addition, these systems may implement MTM and conventionalstructures operating in different frequency ranges or bands, e.g., oneor more frequency bands.

The Doherty architecture can be implemented to include two poweramplifiers that are evenly fed with input power and can be designed tooperate in any of the classes such as class A/B/AB/C/D/E/F/F-1/J. It ispossible to unevenly feed the Doherty PA by applying more power to thepeak PA than to the carrier PA in order to fully modulate the loadimpedance. This results in more output power than the case in which thePAs are driven with the same amount of power.

FIG. 6B illustrates one implementation of the PA system presented inFIG. 6A. This MTM Doherty PA system includes two PAs, 617 and 619,coupled to a 2-way MTM power splitter 613 and a 2-way MTM power combiner615 such as the 2-way MTM power combiners/splitters described in FIGS.3A-3B and FIGS. 4A-4B. The two PAs comprises two power amplifiers: one“main PA” 617 or “carrier PA” and one “auxiliary PA” 619 or “peak PA”.These MTM power combiners/splitters may have a small footprint and lowlosses, and, therefore, can enhance efficiency. In addition, these MTMpower combiner/splitters may also support multi-band frequencies asdescribed in the previous section and integrate input matching network(IMN) and/or output matching network (OMN).

FIG. 6C illustrates another implementation of the PA system presented inFIG. 6A. In FIG. 6C, an N-way MTM Doherty PA system is comprised of an NPAs 629, 631 coupled to an N-way MTM power splitter 623 at its input andan N-way MTM power combiner 625 at its output such as the N-way MTMpower combiners/splitters described in FIGS. 3A-3B and FIGS. 4A-4B. Inthis example, the PAs are comprised of a single main or carrieramplifier 629 and (N−1) peak or auxiliary amplifiers 631. An N-wayDoherty PA may have the ability to operate at a greater back-off whileretaining high efficiency and high linearity. By applying the MTMtechnology to the Doherty PA system, the MTM Doherty PA system can beconstructed in a compact form and have multi-band properties. When thisarchitecture is unevenly driven, the output power of the system may beincreased due to the unevenly driven split. For example, the auxiliaryor (peak) amplifier 631, characterized by a switching amplifier, turnsON only when there is a certain power applied to the input of theswitching amplifier 631. When the input power is low, the switchingamplifier 631 is turned OFF. Therefore, by using unevenly drivensplitters 623, the auxiliary amplifiers 631 can be turned ON earlier orlater than in the evenly driven case. The impedance of each branch ofthe power splitter 623 may be matched by selecting the parameters thatdefines the CRLH and by design to the input impedance of the PA. Thesame is true for the power combiner 625 located at the output. Eachbranch of the power combiner 625 may be matched to the output impedanceof the PAs 629, 631. It is also possible to use a series 90° line at theoutput of each PAs 629-631 as described in FIG. 14. The only differenceis the use of 90° and 270° instead of 0° and 360°.

In yet another implementation of a PA system presented in FIG. 6A, theN-way power splitter/combiner described in FIG. 6A can be replaced witha 2-way MTM Wilkinson splitter/combiner as described in FIG. 5A.However, the design may be extended to N-way splitter/combiners such asthose described in FIGS. 4A-4B and based on the N-way MTM Wilkinsonsplitter/combiners described in FIG. 5B. Examples of various PAarchitectures that may use the 2-way MTM Wilkinson powersplitter/combiners shown in FIG. 5A are illustrated in FIGS. 7-10 andare described herein as follows.

In the MTM Doherty PA system shown in FIG. 7, the input stage of the MTMDoherty PA system may be comprised of a 2-way MTM Wilkinson splitter 701as shown in FIG. 5A and described previously. In this implementation,the Wilkinson splitter may utilize an evenly divided splitter where k=1.Two PAs 705 and 707 are coupled to the output ports of the MTM Wilkinsonsplitter 701 in which the power is evenly divided into the input of amain amplifier 705 and the input of an auxiliary amplifier 707.Alternatively, each PA can be fed with an arbitrary amount of inputpower which can be designed by selectively choosing and splitting theinput signal to the PAs 705, 707. The amplified signals from eachamplifier 705, 707 are fed into a output power combiner 709 in whichconventional RH components (713, and 717) can be used at the outputstage of the PA system.

The MTM Doherty PA system in FIG. 8 utilizes the same splitter 701 andtwo PAs 705 and 707 shown in FIG. 7 and previously described. In thisimplementation, the output power combiner 709 can be replaced with anMTM output power combiner 801 comprised of CRLH-TLs. In contrast to theconventional case, the MTM components at the output stage offer anincreased bandwidth of the Doherty amplifier due to a flatter phaseslope.

The MTM Doherty PA system in FIG. 9 also utilizes the same splitter 701and two PAs 705 and 707 shown in FIG. 7 and previously described. Inthis implementation, the conventional RH components 709 having a phaseof +90 degrees are replaced with MTM quarter wavelength components 901at the output stage. The bandwidth of the inverted Doherty may also beincreased as a result of the MTM components 901 supplied at the outputstage.

In another implementation, the MTM Doherty PA system shown in FIG. 10 iscomprised of the same MTM Wilkinson splitter 701 and two PAs 705 and 707shown in FIG. 7 and previously described. In this configuration, asignal from the main amplifier 705 and another signal from the auxiliaryamplifier 707 are coupled and fed to corresponding input ports of a MTMWilkinson combiner 1001.

One technique to build power amplifiers with enhanced efficiency andlinearity is known as the Envelope Elimination and Restoration (EER).FIG. 11A illustrates an example of an EER system 1100. In the EER system1100, an RF signal at an input port 600 is split into two paths: anupper path 1101 and a lower path 1103. In the upper path 1101, the RFsignal at the input port 600 is fed into an envelope detector 1101 fordetecting an envelope of the input signal. The envelope detector 1105 isin series with and coupled to a pulse width modulator (PWM) 1107 fordigitizing the envelope signal from the detector 1105. The PWM 1107 iscoupled to a Class-S Modulator 1109 which receives and amplifies thedigitized envelope signal transmitted by PWM 1107. A low pass filter(LPF) 1111, which is coupled to the output of the Class-S Modulator1109, is used for recovering the desired frequency range to the biasinput of a conventional PA 1117. In the lower path 1103, the RF signalat the input port 600 is fed into a limiter 1113 for eliminating theamplitude modulation components and retaining the phase portion of thesignal. The limiter 1113 is further coupled to a time delay 1115 forcompensating delay of the modulated signal in the upper path 1101. Thetime compensated signal at the output of the time delay 1115 is fed intothe conventional PA 1117 for amplification to produce an output signalat the output port 607, which is proportional to the RF input signal atthe input port 600.

The EER architecture in FIG. 11A may suffer several disadvantages. Onedisadvantage is switching losses coming from the Class-S modulator 1109which can increase when the frequency increases. Another disadvantage isclipping problems occurring at the pulse width modulator (PWM) 1107caused by signals that have large envelope variations and non-linearityoriginating at the output portion of the envelope detector 1005. Toaddress the large envelope variations, one solution is to replace thepulse width modulator (PWM) 1107, the Class-S modulator 1109, and lowpass filter 1111 with a delta-sigma modulator, which also rejects thenoise outside the band of interest. In addition, by including MTMfilters in the envelope delta-sigma modulated (EDSM) scheme, the bandpass filter can achieve a desired bandwidth having a lower loss thanwith conventional filters. In addition, for this EER architecture, a lowpass MTM filter may be utilized. MTM PAs of class-F, inverse class-F, orclass-J, for example, can include an integrated filter which is part ofthe output matching network (OMN) of the PA. Such design can have theadvantage of reducing losses introduced when connecting to aconventional filter. High efficiency may be obtained by utilizing afilter that is connected at the output of the PA. This type of an MTMfilter design with a harmonic tuner can improve the amplifier'slinearity.

In an exemplary implementation presented in FIGS. 11B-11D, an MTM PA oranother MTM circuit element is integrated in the Envelope EliminationRestoration (EER) to enhance the EER.

For example, the EER system in FIG. 11B illustrates an RF signal at aninput port 600 that is split into two paths: an upper path 1121 and alower path 1123. In the upper path 1121, the RF signal at the input port600 is fed into an envelope detector 1105 for detecting an envelope ofthe input signal. Envelope detector 1105 is in series with and coupledto a delta-sigma modulator 1125 which digitizes the envelope signal fromthe detector 1105. The delta-sigma modulator 1125 is coupled to amodulator 1127 that combines the two paths where the output of thedelta-sigma modulator is recombined with the signal in the lower path1123. In the lower path 1123, the RF signal at the input port 600 is fedinto a limiter 1113 for eliminating the amplitude modulation componentsand retaining only the phase portion of the RF signal. The output of thelimiter 1113 is coupled to a time delay 1115 for compensating the delayof the modulated signal in the upper path 1121 so that the signals inthe two paths are synchronized in phase at the modulator 1127. The timecompensated signal at the output of the time delay 1115 is fed intomodulator 1127 and is recombined with the envelope signal in the upperpath 1121. In this implementation, the output signal at modulator 1127is fed into an MTM PA for amplification to produce an output signal atthe output port 607, which is proportional to the RF input signal at theinput port 600. In another implementation, the MTM PA 1129 can beintegrated with a harmonic tuner and a band pass filter which is part ofthe output matching network of the PA. By integrating the PA with otherMTM components, the overall circuit size of the PA can be reduced whileimproving the performance of the PA.

FIG. 11C illustrates an implementation of the conventional EER system ofFIG. 11A. In this implementation, the amplified RF signal generated atthe output of the PA 1117 can be coupled to an MTM filter 1137. Byincluding an MTM filter in the EER system of FIG. 11A, a desiredbandwidth can be achieved for the band pass filter. In this example, themodulated scheme can be an envelope delta-sigma modulated (EDSM) 1125scheme or PWM/Class-S/LPF modulated scheme (1107, 1109, 1111) as shownin FIG. 11A.

In FIG. 11D, the PA 1117 and MTM filter 1137 of FIG. 11C are replaced byan MTM PA 1141. The MTM PA 1141 is a PA with a CRLH MTM structure toenhance the PA performance and to reduce the PA dimension. The MTM PA1141 may also include an integrated filter. Because the MTM PAs aredesigned as a block, the MTM PA 1141 can be optimized with all the MTMcomponents. This design integration methodology can have severaladvantages, such as the ability to reduce circuit size and improveoverall performance of the PA. The MTM PA can be designed as class-F,inverse class-F, or class-J power amplifiers.

A MTM-based multi-PA system can also include an input port that receivesan input RF signal and multiple transmission lines based on CRLH MTMstructures and connected in series to receive the input RF signal fromthe input port. Each transmission line has a CRLH MTM structure toproduce, at a given operating frequency or resonance, a zero phase delayor a phase delay of n360° where n is a non-zero integer and including abranch output port that splits a portion of the RF signal as arespective branch RF signal for the transmission line and differenttransmission lines producing different respective branch RF signals.This system includes multiple power amplifiers that are coupled to thebranch output ports of the transmission lines, respectively where eachpower amplifier receives and amplifies a respective branch RF signal toproduce an amplified branch RF signal. A power combiner based on a CRLHMTM structure is coupled to receive the amplified branch RF signals fromthe power amplifiers and to combine the received amplified branch RFsignals into a combined output RF signal.

FIG. 12 shows one example of such a system where a series fed parallelcombining distributed amplifier includes zero-degree CRLH TLs 1203. Atthe input of each transistor 1209 there may be a zero degree CRLH TL1203. All of these zero degree CRLH TLs form a series zero degree CRLHTL which is terminated with a load 1205. An MTM power combiner 1207 iscoupled to the outputs of the transistors 1209 to combine the amplifiedoutputs into a single output signal. The zero degree CRLH TL 1203 andthe MTM power combiner 1207 can be designed to provide multi-bandoperations and optimize the matching and power transfer in multiplebands. In addition, the zero degree CRLH TL 1203 provides filtering of2nd and 3rd harmonic frequencies because the characteristic impedancecan be a function of the signal frequency. Therefore, the efficiency ofthe structure shown in FIG. 12 may be enhanced in comparison to someother PA structures.

In yet another implementation shown in FIG. 13, a series fed parallelcombining distributed amplifier may include a zeroth order resonator. Inthis implementation, there may be a zeroth order resonator at the input600 which provides a stationary wave at the input in the resonator witha filtering effect and no energy dissipated at the load 1205 as in theprevious case. As a result, a better efficiency and a better linearitymay be obtained.

The exemplary implementations illustrated in FIGS. 14 and 15 depict thezero degree CRLH TL and the zeroth order resonator used in FIGS. 12 and13, respectively. Under these designs, a dual-band or multi-band mainCRLH transmission line is structured to exhibit, at least, a first phaseat a first signal frequency f1 and a second phase at a second, differentsignal frequency f2. This main CRLH transmission line includes CRLH unitcells coupled in series and each CRLH unit cell has a first electricallength that is a multiple of +/−180 degrees at the first signalfrequency and a second, different electrical length that is a differentmultiple of +/−180 degrees at the second signal frequency. Two or morebranch CRLH feed lines are connected at different locations on the CRLHtransmission line to combine signals in the CRLH feed lines into theCRLH transmission line or to divide a signal in the CRLH transmissionline into different signals to the CRLH feed lines. Each branch CRLHfeed line includes at least one CRLH unit cell that exhibits a thirdelectrical length that is an odd multiple of +/−90 degrees at the firstsignal frequency and a fourth, different electrical length that is adifferent odd multiple of +/−90 degrees at the second signal frequency.As illustrated, each CRLH feed line 1020 is connected to a locationbetween two adjacent CRLH unit cells or at one side of a CRLH unit cell.

The device in FIG. 14 includes a dual-band main CRLH transmission line1410 with dual-band CRLH TL unit cells 1412 and branch CRLH feed lines1420. Each unit cell 1412 is designed to have an electrical signallength equal to a phase of zero degree at the first signal frequency f1and a second electrical signal length equal to a phase of 360 degrees atthe second signal frequency f2. Each branch CRLH feed line 1420 includesone or more CRLH unit cells and is configured as a dualband CRLH TLquarter wavelength transformer. The optimum impedances are transformedvia the CRLH TL quarter wavelength transformer 1420 of a length L at 2different frequencies, f1 and f2. In this particular example, each CRLHfeed line 1420 is designed to have a phase of 90° (λ/4) [modulo π] atthe first signal frequency f1 and a phase of 270° (3λ/4) [modulo π] atthe second signal frequency f2. This device has 0 degree phasedifference at one frequency and 360° at another frequency between eachport.

The device in FIG. 15 illustrates a dual-band resonator serial powercombiner/divider. This dual-band CRLH TL performs as a resonator bybeing terminated with an open ended. The output/input ports (port 1-N)can be matched to 50Ω, while the other ports are match to optimumimpedances. These optimum impedances are transformed via a CRLH TLquarter wavelength transformer of length L at 2 different frequencies,f1 and f2. By way of example f1 has a phase of 90° (λ/4) [modulo π]while f2 has a phase of 270° (3λ/4) [modulo π].

The two signal frequencies f1 has f2 do not have a harmonic frequencyrelationship with each other. This feature can be used to comply withfrequencies used in various standards such as the 2.4 GHz band and the5.8 GHz in the Wi-Fi applications. In this configuration, the portposition and the port number along the dual-band CRLH TL 1410 can beselected as desired because of the zero degree spacing at f1 and 360° atf2 between each port.

In another implementation shown in FIG. 16, an MTM amplifying antenna iscomprised of a serial zeroth order resonator power divider 1601 at theinput of each transistor 1605 and a zeroth order resonating antenna 1603at the output of each transistor 1605. Therefore, in this architecture,the output signals are combined in the zeroth order resonating antennawithout a power combiner. U.S. Pat. Nos. 7,482,893 and 7,330,090disclose examples of zeroth order resonating antennas and power dividerswhich can be used to implementing the design in FIG. 16 and areincorporated by reference as part of the disclosure of this document. Inother implementations, the antenna 1603 may be implemented by a leakywave antenna.

The MTM technology can be used to provide broadband or multi-band powercombiner/splitters in compact packages. Some implementations can use3-dB power combiner/splitters to decrease losses in the MTM balanced PA,thereby improving the overall efficiency of the system.

In the example shown in FIG. 17, a balanced PA is comprised of aquadrature coupler such as an MTM branch line coupler 1701, twoidentical PAs (1702, 1703), and an MTM branch line coupler 1705. The RFsignal is received at the input port 600 of the MTM branch line coupler1701 and split equally into two outputs 1711 having a 90° phasedifference. The signals at two outputs 1711 are fed into PA1 1702 andPA2 1703. The remaining port of the MTM branch line coupler 1701 isterminated with a resistor R 1707. The outputs of the PA1 1702 and PA21703 are connected to two inputs 1713 of the MTM branch line coupler1705. One output of the MTM branch line coupler 1705 is used as the RFoutput 607 and the other one is terminated by using a resistor R 1709.

The advantage of using the quadrature coupler at the input is that thereflected signals caused by the impedance mismatch at inputs of the PAscan return to the RF input port in anti-phase. Therefore, the reflectedsignals are canceled, and the input voltage standing wave ratio (VSWR)may be better than the one without the quadrature coupler.

The same reason generally applies to the usage of a quadrature couplerat the outputs of the PAs. The output VSWR can be improved by usingother quadrature couplers such as, for example, a microwave branch linecoupler or a Lange coupler. However, the conventional microwave branchline coupler only provides a narrow band and is generally bulky, whilethe Lange coupler requires complicated fabrication processes such aswire bonding or microwave bridge.

The PA systems can also be designed to support either a wide operationalbandwidth or at least two bands.

In one example shown in FIG. 18, a multi-band MTM branch line couplerincludes four CRLH-TL branches, where two ends of the CRLH-TL1 1801,1803 are connected to the CRLH-TL2 1805, 1807 and two ends of theCRLH-TL2 1805, 1807 are connected to the CRLH-TL1 1801, 1803. CRLH-TL11801, 1803 has phases of θ1 at f1 and θ2 at f2 while CRLH-TL2 1805, 1807has phases of θ3 at f1 and θ4 at f2, where f2 is larger than f1. The θ1,θ2, θ3 and θ4 have to satisfy the phase requirement listed in FIG. 18 inorder to provide a 3 dB power division and quadrature phase between twooutputs. In addition, the characteristic impedance of CRLH-TL1 1801,1803 and CRLH-TL2 1805, 1807 are 35.35Ω and 50Ω, respectively. By usingMTM technology, these branch line couplers can be made smaller in sizeas compared to the conventional branch line couplers.

FIG. 19 illustrates an exemplary dispersion diagram of a CRLH-TL, whichmay be a nonlinear curve and may have zero phase delay at a non-DCfrequency. As a result, the phase advance/delay of θ1 and θ2 can bearbitrarily chosen at arbitrary frequencies of f1 and f2.

PA systems with a wide operational bandwidth can use a branch linecoupler with a wide bandwidth. FIG. 20A shows an example where abroadband branch line coupler includes four CRLH-TLs 2011, 2013, 2015,2017 coupled to form a 4-port coupler. In one implementation, P1 2001 isthe input and P2 2002 and P3 2003 are the two outputs, and P4 2004 canbe designed to handle reflection or to be isolated from the other ports.Due to the symmetry of the branch line coupler, the input and outputports can be used interchangeably. By choosing θ1=−90° and θ2=+90°, thebranch line coupler may provide broader bandwidth as compared to theconventional branch line coupler.

FIG. 20B illustrates a comparison of the return loss and insertion lossbetween the broadband MTM branch line coupler and conventional branchline coupler. In comparison to the conventional case, the MTM branchline coupler covers a broader frequency range in both return loss andinsertion loss responses. FIG. 20C also affirms that the bandwidth ofthe phase difference between |S21| and |S31| is wider for the MTM branchline coupler than the conventional branch line coupler.

FIG. 21 illustrates another implementation of the MTM broadband couplerknown as a planar MTM forward-wave (FW) directional coupler. Examplesfor such an MTM broadband coupler re described in the U.S. patentapplication Ser. No. 12/340,657 which is incorporated by reference aspart of this document. This planar MTM FW directional coupler isimplemented by cascading two coupled metamaterial cells horizontally.The coupled metamaterial cell is realized by paralleling twometamaterial unit cells, for example, 2100-1 and 2100-2, with anadditional inductor L_(ml) connecting between the two cells. Eachmetamaterial unit cell 2100 comprises two transmission lines, two seriescapacitors of 2C_(L) and one shunt inductor of L_(L). In this exemplaryimplementation shown in FIG. 21, the two transmission lines associatedwith an input port, Port 1 2101, are represented by 2105 and 2107. Thetwo transmission lines associated with a transmission port, Port 2 2102,are represented by 2115 and 2117. The two transmission lines associatedwith a coupling port, Port 4 2104, are represented by 2111 and 2113. Thetwo transmission lines associated with an isolation port, Port 3 2103,are represented by 2109 and 2119. The mutual inductor and capacitor ofL_(m) and C_(m) are introduced when two transmission lines of each unitcell are close to each other. See transmission lines 2107 and 2109, forexample. These two transmission lines may possess the intrinsic seriesinductance of L_(R) along the line and the shunt capacitance C_(R)between the lines and the ground. This planar MTM FW directional couplercan provide a wide band of return loss and 3 dB insertion loss at eachoutput port. The planar MTM FW directional coupler described above canbe implemented in the balanced power amplifier for the input and outputsections.

FIGS. 22A-22B illustrates another implementation of the MTM couplerknown as a vertical MTM FW directional coupler. Examples of thisvertical MTM FW directional coupler are also described in the U.S.patent application Ser. No. 12/340,657. The FW coupler in thisimplementation may be constructed by cascading two coupled metamaterialcells. In this exemplary implementation, the coupled metamaterial cellis built by paralleling two metamaterial transmission lines 2201, 2203vertically with an additional inductor L_(ml) 2231 connecting betweenthe two metamaterial transmission lines which indicates one metamaterialtransmission line 2201 is on top of the substrate and the othermetamaterial transmission line 2203 is on the bottom of the substrate.The cross-sectional view of this MTM FW coupler is shown in FIG. 22B.Each metamaterial transmission line 2201, 2203 includes two metamaterialunit cells. Each unit cell comprises two sections of transmission line,two series capacitors of 2C_(L) and one shunt inductor of L_(L). Thevertically coupled transmission lines provide mutual inductance L_(m)and mutual capacitance C_(m). In addition, each port of the metamaterialcoupled lines is connected to the 50Ω CPW 2221 through a CPW bend. Inthis implementation, there are a total of four CPW lines 2221 and fourCPW bends 2209. The MTM FW coupler in this example may also provide abroad bandwidth of return loss and 3 dB insertion loss at each outputport. The vertical MTM FW directional coupler can be used in thebalanced power amplifier for the input and output sections.

Linear amplification using non-linear components (LINC) is formed byadding two signals vectorialy to achieve power amplification. In theexemplary implementation presented in FIG. 23, a LINC architectureincludes MTM LINC (non-linear) PAs 2305 coupled to an N-way MTM PowerSplitter 2301 at the input stage and N-way MTM Power Combiner 2303 atthe output stage. Alternatively, conventional LINC PAs can be used. MTMPAs can support multi-band RF signals with its compact size and providelow loss, resulting in high efficiency. One disadvantage of conventionalLINC amplifiers is the complexity of the hardware; some applicationsrequire extensive signal processing, which makes the LINC architecturebulky and expensive. The MTM PAs may reduce the complexity of thehardware and provide broadband with high efficiency. The MTM can achievea flat slope for the phase in a certain range of frequencies. In otherwords, the same phase can be achieved at different frequencies which canimprove the phase matching of the combined PAs.

In an exemplary implementation of a phase combining system, shown inFIG. 24, different MTM non-linear PAs 2403 may be fed with differentsignals and different phases, which may be combined in phase with anN-way MTM power combiner 2405 or other power combiners such as an MTMWilkinson power combiner shown in FIG. 5A-5B and previously described.Each PA is fed with an RF input signal. In this configuration, the useof MTM structures can be designed to allow a certain frequency to passwhile blocking other frequencies. Therefore, the PA architecture of FIG.24 can implement this design when input signals have different frequencybands or a single frequency band.

While this document contains many specifics, these should not beconstrued as limitations on the scope of any invention or of what may beclaimed, but rather as descriptions of features specific to particularembodiments. Certain features that are described in this document in thecontext of separate embodiments can also be implemented in combinationin a single embodiment. Conversely, various features that are describedin the context of a single embodiment can also be implemented inmultiple embodiments separately or in any suitable subcombination.Moreover, although features may be described above as acting in certaincombinations and even initially claimed as such, one or more featuresfrom a claimed combination can in some cases be excised from thecombination, and the claimed combination may be directed to asubcombination or variation of a subcombination.

Thus, particular implementations have been described. Variations andenhancements of the described implementations and other implementationscan be made based on what is described and illustrated.

1. A device based on a Composite Right and Left Handed (CRLH)Metamaterial (MTM) structure, comprising: a first transmission lineincluding a Composite Right and Left Handed (CRLH) Metamaterial (MTM)structure, a first terminal that is terminated at a first impedance anda second terminal; a plurality of second transmission lines of CRLH MTMstructures that are coupled to the second terminal of the firsttransmission line and are configured to either split an input RF signalreceived from the second terminal of the first transmission line into aplurality of output RF signals in the second transmission lines,respectively, or to combine a plurality of input RF signals in thesecond transmission lines into an output RF signal at the secondterminal of the first transmission line, each second transmission linehaving a first terminal that is coupled to the second terminal of thefirst transmission line and a second terminal that is terminated at arespective second impedance, wherein the first transmission line and thesecond transmission lines are structured to provide impedance matchbetween the first impedance at the first terminal of the firsttransmission line and a respective second impedance of each of thesecond transmission lines, and wherein the first transmission line andthe second transmission lines are structured to be in a firstconfiguration where the first transmission line is a CRLH MTMtransmission line of an electrical length having a phase delay of ±m90°where m is a positive odd integer, and each of the second transmissionlines is a CRLH MTM transmission line of an electrical length having azero phase delay or a phase delay of ±n180° where n is a positiveinteger, or in a second configuration where the first transmission lineis a CRLH MTM transmission line of an electrical length having a zerophase delay or a phase delay of ±m180° where m is a positive integer andeach of the second transmission lines is a CRLH MTM transmission linehaving a phase delay of ±n90° where n is a positive odd integer.
 2. Thedevice as in claim 1, wherein: the second transmission lines havedifferent impedances.
 3. The device as in claim 1, wherein: the secondtransmission lines have substantially equal impedances.
 4. The device asin claim 1, wherein: the first transmission line has resonances at twoor more different RF frequencies; and each second transmission line isstructured to have resonances at the two or more different RFfrequencies of the first transmission line.
 5. A power amplifier system,comprising: an RF input port configured to receive an input RF signal;an envelope detector coupled to the RF input port and configured toreceive a first portion of the input RF signal and to extractinformation on an amplitude envelope of the input RF signal to producean envelope signal; a sigma-delta modulator coupled to the envelopedetector and configured to convert the envelope signal into a firstsignal; a limiter circuit coupled to the RF input port and configured toreceive a second portion of the input RF signal and to eliminateamplitude modulation components of the input RF signal to produce asecond signal that retains phase of the input RF signal; a time delaycircuit coupled to the limiter circuit and configured to cause a delayin the second amplifier input signal to synchronize with the firstsignal; a modulator coupled to receive the first and second signals andconfigured to combine the received first and second signals into a thirdsignal; and a power amplifier coupled to the modulator and configured toreceive and to amplify the third signal into an output RF signal that isproportional to the input RF signal, wherein the power amplifier is aComposite Right and Left Handed (CRLH) Metamaterial (MTM) structurepower amplifier.
 6. A power amplifier system, comprising: an RF inputport configured to receive an input RF signal; an envelope detectorcoupled to the RF input port and configured to receive a first portionof the input RF signal and to extract information on an amplitudeenvelope of the input RF signal to produce an envelope signal; asigma-delta modulator coupled to the envelope detector and configured toconvert the envelope signal into a first signal; a limiter circuitcoupled to the RF input port and configured to receive a second portionof the input RF signal and to eliminate amplitude modulation componentsof the input RF signal to produce a second signal that retains phase ofthe input RF signal; a time delay circuit coupled to the limiter circuitand configured to cause a delay in the second amplifier input signal tosynchronize with the first signal; a power amplifier coupled to receivethe first and second signals and configured to produce an output RFsignal that is proportional to the input RF signal; and a filter coupledto the power amplifier and configured to filter the output RF signal andhaving a Composite Right and Left Handed (CRLH) Metamaterial (MTM)structure.
 7. The system as in claim 6, wherein: the power amplifier isa Composite Right and Left Handed (CRLH) Metamaterial (MTM) poweramplifier.
 8. A power amplifier system for amplifying an RF signal,comprising: a Zero^(th) order resonator power divider having a CompositeRight and Left Handed (CRLH) Metamaterial (MTM) structure and configuredto receive an input RF signal and to split the received input RF signalinto branch RF signals in phase with respective to one another; aplurality of transistors coupled to the Zero^(th) order resonator powerdivider and configured to receive the branch RF signals, respectively,and to amplify the received branch RF signals; and a Zero^(th) orderresonating antenna based on a CRLH MTM structure and coupled totransistors to receive amplified branch RF signals.
 9. A power amplifiersystem, comprising: an RF input configured to receive an input RFsignal; a first quadrature coupler based on a Composite Right and LeftHanded (CRLH) Metamaterial (MTM) structure and having a first input portcoupled to the RF input, the first input port being configured toreceive the input RF signal, a second input port that is terminated, afirst output port and a second output port, the first quadrature couplerbeing configured to split splitting the input RF signal into a first RFsignal at the first output port and a second RF signal at the secondoutput port, the second RF signal having a phase shift of 90 degreeswith respect to the first RF signal; a first power amplifier coupled tothe first output port and configured to receive and to amplify the firstRF signal; a second power amplifier coupled to the second output portand configured to receive and to amplify the second RF signal; and asecond quadrature coupler based on a CRLH MTM structure and having afirst input port coupled to receive the amplified first RF signal fromthe first power amplifier, a second input port coupled to receive theamplified second RF signal from the second power amplifier, a firstoutput port that is terminated and a second output configured to outputan RF output signal, the second quadrature coupler configured to combinesignals received at the first and second input ports with a 90-degreephase shift to produce the RF output signal.
 10. The power amplifiersystem as in claim 9, wherein: one of the first and second quadraturecouplers is a CRLH MTM branch line coupler formed by four CRLH MTMtransmission lines.
 11. The power amplifier system as in claim 10,wherein: the first and second quadrature couplers have CRLH MTMstructures with resonances at two or more RF frequencies.
 12. The poweramplifier system as in claim 9, wherein: one of the first or secondquadrature couplers is a planar forward-wave directional coupler basedon a CRLH MTM structure that includes unit cells formed in commonmetallization layers at different horizontal locations in a plane of asubstrate in which the unit cells are formed.
 13. The power amplifiersystem as in claim 9, wherein: one of the first or second quadraturecouplers is a vertical forward-wave directional coupler based on a CRLHMTM structure that includes unit cells formed in different metallizationlayers along a vertical direction perpendicular to a place of asubstrate in which the unit cells are formed.
 14. An RF communicationapparatus comprising: a system configured to produce an RF signal; andthe power amplifier system according to claim 5, coupled to receive theRF signal at the RF input port.
 15. An RF transmission systemcomprising: a modulation system configured to produce an RF signal; andthe power amplifier system according to claim 6, coupled to receive theRF signal at the RF input port.
 16. An RF transmission systemcomprising: a modulation system configured to produce an RF signal; andthe power amplifier system according to claim 8, coupled to receive theRF signal at the RF input port.
 17. An RF transmission systemcomprising: a modulation system configured to produce an RF signal; andthe power amplifier system according to claim 9, coupled to receive theRF signal at the RF input port.